Transceiver architectures rely on frequency conversion using local oscillator to generate either radio frequency (RF) or intermediate frequency (IF) local signals to up-convert baseband signals to RF signals or vice versa. So, the oscillator is one of the most important components in each wireless system. The local oscillator performances affect wireless systems as described in the following.
Phase noise is one of the most critical impairments of radio systems, as it corrupts the information carried in the phase of the carrier due to its non-ideality. There are two different types of corruptions that phase noise can make in the systems. One is from in-band, i.e. modulated signal band phase noise, which directly affects down converted or up converted signals. The amount of in-band phase noise, other than the one that is very close to the carrier, which is slow varying enough for the data to be affected, can be represented as a phase error or error vector magnitude (EVM).
Another key aspect of phase noise impact can be explained with an interferer, which is often called a blocker, is shown in FIG. 1. A mixer 100 is mixing a radio frequency signal RF comprising a desired signal 102 and an interfering signal 104 with a local oscillator signal LO having a local oscillator frequency spectrum 110. The mixer 100 produces an IF signal which frequency spectrum is obtained by a superposition of the local oscillator frequency spectrum 110 mixed by the desired signal 102 and the local oscillator frequency spectrum 110 mixed by the interfering signal 104. When a blocker gets up or down converted, the LO's phase noise will override its phase noise onto the blocker and parts of the phase noise will stay on top of the signal band. These types of non-ideal phase noise can impact any type of radio system such as wideband code division multiple access (W-CDMA) and global system for mobile communications (GSM). The destructive effect of phase noise can be best seen in the front end of a super-heterodyne receiver.
Since orthogonal frequency division modulation (OFDM) signals transmit multi-carriers in the frequency domain, multi-carriers with the same phase noise on each carrier can be analyzed as one single-carrier with the same phase noise. However, the far-out phase noise can cause inter symbol interference (ISI), which means there is going to be a system requirement on phase noise for this system as well. Long-term Evolution (LTE) and 802.11g systems use OFDM techniques.
On the other hand, the frequency spectrum is a valuable asset in the wireless communications. The number of wireless users demands more efficient usage of frequency resources. The communication transceivers rely on frequency conversion using local oscillator and therefore the spectral purity of the oscillators in both the receiver and transmitter is one of the factors limiting the performance and maximum number of available channels and users.
Conventionally, cross coupled negative, i.e. −gm, oscillators usually form the basis for low noise high performance oscillator designs. Moreover, tail-current shaping, operation in class-C mode and higher oscillators may also be deployed. However, the known oscillators suffer from the fact that most circuit noise converts to phase noise. This limits the lowest phase noise achievable. We review the status of the latest prior technologies.
FIG. 2A shows a schematic diagram of a clip and restore (C&R) oscillator (DCO) or voltage controlled oscillator (VCO) as described by A. Visweswaran, R. B. Staszewski, J. R. Long: “A clip and restore technique for phase desensitization in 1.2 V 65 nm CMOS oscillator for cellular mobile and base stations”. Positive feedback is realized between the drain and gate terminals of transistors M1 and M2 using 1:2 step-up transformers T1 and T2. The transistors M1 and M2 are realized as thick oxide devices withstanding to large swing. Large swing decreases rise and fall time of output oscillation voltage and ensures hard clipping at output. Separate transformers T1 and T2 provide common mode rejection, adequate coupling, that means not too week, and are designed to control impedance transformation at their interface. The voltage gain is realized by winding step-up and impedance mismatch between drain and gate by the transformer turns ratios and coupling factor. As signal amplitude 201 at gate decreases, the transistors M1 and M2 transition quickly from saturation 206 to triode 212, causing output signal 203 to clip as can be seen from FIG. 2B. The transition regions 204 are small. The bias voltage (VB) is varied to adjust the onset of clipping. The fundamental tone excites the secondary winding of T1, while harmonics are shunted to ground through capacitance CF2. The transistor gates are driven by the second transformer T2, which steps up the filtered waveform for peak-to-peak swing more than the traditional oscillators. This strong overdrive ensures hard clipping at the transistor output. According to the linear time-variant phase noise model, the impulse sensitivity function quantifies phase noise sensitivity across one period of oscillation. Note that drain voltage 203 is clipped and the ISF value is zero during the clipping time 202. Hence, noise injected by the passive and active devices as well as power supply does not perturb the oscillator zero-crossings or phase in the clipping time. Capacitance CF1 improves frequency pushing and decreases sensitivity of output frequency to P-type metal-oxide-semiconductor logic (PMOS) parasitic. Switched capacitor circuit CX1 is used for course tuning the tank while switched capacitor circuit CX2 is used for fine tuning the tank. Node N is used for adjusting the drain voltage clipping and current consumption of the oscillator.
The document of T. Lee, A. Hajimiri: “Oscillator Phase Noise: A tutorial,” in IEEE J. Solid State Circuits, Vol. 35, no. 2, pp. 326-336, March 2000 describes that signal amplitude and resonator quality factor should be maximized to get a better phase noise performance in the 20 dB/Dec section. The Impulse Sensitivity Function (ISF) specifies the contribution of the circuit noise components to the phase noise in terms of waveform properties of the oscillator. The direct current (DC) value of the effective ISF should be made as close to zero as possible to suppress the up-conversion of 1/f noise.
The document of Mazzanti and P. Andreani: “Class C Harmonic CMOS VCOs, With General Result on Phase Noise” in IEEE J. Solid State Circuits, vol. 43, NO. 12, pp. 2716-2729, December 2008 describes that a large tail capacitance at the common source node of the differential pair, together with a bias network to prevent the transistors from working in the deep triode region, allows at the same time a high conversion efficiency of the DC bias current in to the fundamental current component of gm device current and a minimum generation of the effective noise of the tank, thus maximizing the figure of merit performance.
The document of E. Hegazi, H. Sjoland, A. Abidi: “A Filtering Technique to lower Oscillator Phase Noise,” in ISSCC Dig. Tech. papers, February 2001 describes a capacitor and an inductor, in series with a current source forming a low pass filter which shunts noise at 2f from the current source to ground. The additional resonator also provides a high impedance path between the tank and ground. Consequently, the structure doesn't degrade Q of the tank. It can improve phase noise and figure of merit (FoM) of the oscillator.
Base station receiver (RX) oscillator phase noise requirements between 600 kilohertz (KHz) and 3 MHz are extremely difficult to satisfy using a fully monolithic VCO or DCO fabricated in bulk Complementary metal-oxide-semiconductor (CMOS) technology. The GSM -900-BTS and the DCS-1800-BTS RX specifications at 800 kHz of −147 dBc/Hz and −138Bc/Hz, respectively, are considered the most difficult phase noise specifications to meet. The following equation (1) gives the phase noise spectrum of an arbitrary oscillator in the 1/f2 region of the phase noise spectrum.
                                          L            ⁡                          (              Δω              )                                =                      10            ⁢                                                  ⁢                                          log                10                            (                                                                    ∑                    i                                    ⁢                                                                          ⁢                                                            Γ                      rms                      2                                        ⁡                                          (                                                                                                    i                                                          n                              ,                              i                                                        2                                                    /                          Δ                                                ⁢                                                                                                  ⁢                        f                                            )                                                                                        2                  ⁢                                                                          ⁢                                      C                    2                                    ⁢                                                                                    A                        2                                            ⁡                                              (                        Δω                        )                                                              2                                                              )                                      ,                            (        1        )            where in2/Δf is the equivalent noise current mean square density per bandwidth, A is the voltage amplitude of oscillation voltage, Γrms is the root mean square (rms) value of the ISF for the noise current source, C is the tuning capacitance of the resonator tank, ω0 is carrier frequency and (Δω) is the frequency offset from the carrier.
Γrms, the rms value of the impulse sensitivity function, is a dimensionless, frequency and amplitude independent periodic function with a period of 2π which describes how much phase shift results from applying a unit impulse noise at the certain point of oscillation voltage. This parameter is constant for a given oscillator architecture and it is independent from the technology.
Equation (1) illustrates that phase noise improves as both the carrier power and equivalent quality factor Q increase at a given frequency offset. But unfortunately, the oscillation amplitude swing is limited to twice of the voltage source (VDD) in the traditional oscillator structures, and VDD scales down in the advanced CMOS technology. Consequently, the amplitude of oscillation decreases and degrades the phase noise performance. The quality factor of the tank is determined by the quality factor of the inductor of the oscillator. The quality factor will be improved when reducing the DC resistance, skin effect and substrate loss of the inductor. In general, the quality factor of the inductor is limited to 15-30 due to limited width and thickness of the metal line and low resistivity material of the substrate. The maximum allowable width and conductivity of the metal improves a little in newer CMOS technology, but on the other hand the inductor will be closer to lossy substrate and it can reduce the quality factor of the inductor. Consequently, no significant phase noise improvement can be seen in newer CMOS technology.